© H. Nakano et al., published by EDP Sciences, 2019

1 Introduction

A loop antenna radiates a linearly polarized (LP) wave [1]. When the circumference of the loop is one wavelength, the maximum LP radiation appears in the broadside direction. This broadside LP radiation can be changed to circularly polarized (CP) radiation by adding perturbation elements [2] to the loop, as shown in Figure 1. The rotational sense of the CP radiation, either a right-handed (RH) sense or a left-handed (LH) sense, is uniquely determined by the location of the perturbation elements relative to feed point F: the loop in Figure 1 radiates an RHCP wave. In other words, the loop antenna radiates a CP wave with a single rotational sense.

The single rotational sense also holds true for a grid loop array in [3]. Although perturbation elements are not added to the loops, a traveling current is generated along the arrayed loops and radiates a CP wave. The rotational sense is determined by the winding sense of the loops.

It is often required that an antenna has dual-band counter CP radiation to keep sufficient separation of signals, i.e., the LHCP and RHCP radiation in two different frequency bands, in order to avoid interfering with each other. When a pair of dual-band counter CP antennas is adopted as a transmitting antenna and a receiving antenna, it is desirable for these antennas to have the same gain. If the LHCP and RHCP gains are different, the receiving antenna captures LHCP and RHCP waves with different receiving power levels. Therefore, a post-process circuit connected to the receiving antenna needs additional amplification circuits to enhance the weak power. This complicates the post-process circuit designs. To avoid such an issue, the same gain (balanced gain) is desired.

Recent study has shown that a metamaterial (MTM) loop antenna, referred to as a metaloop, is an antenna that meets the requirement of dual-band counter CP radiation [4], where the loop is realized using the concept of a composite RH and LH transmission line [5–7], referred to as a metaline.

The maximum gain for the metaloop within the low frequency band (LoFB) in [4] is smaller than that within the high frequency band (HiFB). This is attributed to the antenna size relative to the free-space wavelength (electrical antenna size); the free-space wavelength within the LoFB is larger than that within the HiFB, and hence the electrical antenna size within the LoFB is smaller than that within the HiFB.

Thus, a question arises as to how the difference in the gains can be reduced. This paper presents a technique for reducing the band gain difference for a dual-band counter CP metaloop antenna, where a parasitic loop (paraloop) is placed above the metaloop. Note that the metaloop antenna in this paper is analyzed using a full wave analysis tool, HFSS [8], and experimental work is performed in an anechoic chamber.

So far, other authors have designed MTM-loaded and MTM-inspired loop antennas. For example, the MTM-loaded loop in reference [9] and MTM-inspired loop [10] are designed as an LP antenna. An antenna using complementary capacitively loaded loop in reference [11] and loop antennas in [12–20] are also designed as an LP antenna. The design requirements for these antennas differ from ours, i.e., dual-band counter CP radiation with balanced gain, while meeting two additional requirements: (1) broadside radiation and (2) simple feed system without balun circuits. To our best knowledge, there have not been such MTM-related loop antennas.

Fig. 1 Natural loop antennas radiating a circularly polarized wave with a single feed point F. The loop has perturbation elements [2].

2 Metaloop structure

The metaloop antenna to be considered here is shown in Figure 2. The antenna arm is printed on a square dielectric substrate of side length S_sub, relative permittivity ε_r and thickness B. The substrate is backed by a square ground plane (GP) of side length S_GP (=S_sub). The antenna arm, symmetric with respect to the x–z plane, is composed of five straight metalines, whose lengths are L₁ = L₅ and L₂ = L₃  = L₄. Each metaline is made of numerous conducting subwavelength segments of width w and length p₀, where the gap between neighboring segments is denoted as Δg. Repeating arm sections, each having length 2(Δg + p₀) ≡ p, are designated as the arm unit cells. The central segment of the cell is short-circuited to the GP through a chip inductor, L_Y. Neighboring segments are connected through a chip capacitor, 2C_Z. Point F is the feed point and point T is the terminal point connected to the GP through a resistive load, R_B, to suppress reflected currents from point T to point F.

Note that L_Y and 2C_Z are determined as follows. First, using HFSS [8], we obtain the frequency response of the scattering parameters (S-parameters) for the unit cell including L_Y and 2C_Z. Second, based on the obtained S-parameters, we draw the dispersion curve (phase constant β = ±2π/λ_g vs. frequency f [4], where λ_g is the guided wavelength). These two steps are repeated until the dispersion curve is balanced (smoothly connected at a preselected transition frequency f_T), changing L_Y and 2C_Z. The values for L_Y and 2C_Z when the dispersion curve is balanced are what we need.

The parameters used for the following discussion are summarized in Table 1; these parameters create a preselected transition frequency of f_T = 3 GHz, i.e., phase constant β is negative at frequencies below f_T and positive at frequencies above f_T. Figure 3 shows the loop peripheral length L₁ + L₂ + L₃ + L₄ + L₅ normalized to the guided wave length, λ_g, as a function of frequency f. Note that the frequency-dependent guided wavelength λ_g used in Figure 3 is derived from the dispersion curve explained in the previous paragraph.

Frequencies f_L and f_U in Figure 3 are the lower- and upper-edge frequencies for a fast wave region. The radiation occurs between f_L and f_U. A traveling wave current flows along the loop from point F to point T, with guided wave length λ_g, which varies with frequency f. The propagation phase constant is β = −2π/λ_g < 0 at f < f_T, β = 0 at f = f_T, and β = 2π/λ_g > 0 at f > f_T. In order to obtain a broadside beam at two different frequencies where the loop length is 1λ_g, the in-phase condition (β = 0) is not used.

Fig. 2 Metaloop antenna. (a) Perspective view. (b) Side view. (c) Arm unit cell.

Fig. 3 Loop length L1 + L2 + L3 + L4 + L5 normalized to guided wavelength λg, as a function of frequency. β < 0 at f < fT, β = 0 at fT, and β > 0 at f > fT.

3 Frequency response of the gain

By resistive load R_B, a traveling wave current flows along the loop from point F to point T, with frequency-dependent guided wavelength λ_g. In other words, the metaline acts as a leaky wave antenna. Generally, the radiation efficiency of the leaky wave antenna is not high due to absorption of the power input to the antenna by R_B. If R_B is removed, the input impedance (and hence Voltage Standing Wave Ratio (VSWR)) becomes unstable and CP radiation is not obtained by a reflected current from point T.

The current at frequency f < f_T has a progressive phase distribution from point F to point T due to a negative phase constant (β = −2π/λ_g < 0). Hence, the current behaves as if it travels from point T to point F (clockwise). This results in LHCP radiation in the broadside direction at a frequency where the loop length is one guided wave length (1λ_g). Conversely, the current at f > f_T has a regressive phase distribution from point F to point T due to a positive phase constant (β = 2π/λ_g > 0). Hence, the current flows from point F to point T (counter clockwise), resulting in RHCP radiation in the broadside direction at a frequency satisfying a 1λ_g-loop length. Thus, dual-band counter CP radiation in the broadside direction is obtained.

From Figure 3, it is expected that CP broadside radiation will be obtained around frequencies 2.60 GHz ≡ f_N and 3.50 GHz ≡ f_H, because the loop length normalized to guided wavelength λ_g is one at f_N and f_H: (L₁ + L₂ +…+L₅)/λ_g = 1. For the remainder of the paper, f_N and f_H are referred to as the Nion frequency and Hion frequency, respectively. Figure 4 shows the gain as a function of frequency, where G_L denotes the gain for an LHCP wave, called the LHCP gain, and G_R denotes the gain for an RHCP wave, called the RHCP gain. It is found that G_L is dominant at frequencies below f_T = 3 GHz, because the current flows with a negative phase constant (β < 0). Conversely, G_R is dominant at frequencies above transition frequency f_T due to β > 0. Maximum gain G_L appears at a frequency near Nion frequency f_N and maximum gain G_R appears at a frequency near Hion frequency f_H. These frequencies are denoted as f_GLmax and f_GRmax, respectively. The difference in the maximum gains, G_Lmax at 2.58 GHz ≡ f_GLmax (close to f_N) and G_Rmax at 3.51 GHz ≡ f_GRmax (close to f_H), is approximately 5.5 dB.

Fig. 4 Frequency response of the gain.

4 Reduction in the gain difference

In this section, the difference in maximum gains G_Lmax and G_Rmax is reduced as much as possible. For this, the antenna system shown in Figure 5 is considered, where a parasitic natural conducting strip loop of width w_para and peripheral length L_para is located at height H_para above the metaloop. The parasitic loop is designed such that only the smaller gain, G_Lmax at f_GLmax, is increased, without reducing the larger gain, G_Rmax at f_GRmax.

The CP radiation from the metaloop excites the parasitic loop and generates a rotating/traveling current along the loop.

The parasitic loop acts as a director for the driven meta loop, like Yagi–Uda antenna. If the radiation from the para sitic loop at f_GLmax is constructively superimposed onto the radiation from the metaloop in the broadside direction, gain G_Lmax increases. For this to occur, parasitic loop length L_para is chosen to be one free-space wavelength (1λ₀) at f_GLmax: L_para = 116.7 mm. The remaining task is to optimize loop height H_para.

Figure 6 shows G_L at f_GLmax = 2.58 GHz and G_R at f_GRmax = 3.51 GHz in the broadside direction (z-direction) as a function of parasitic loop height H_para. It is found that there is an optimum antenna height for G_Lmax ≈ G_Rmax.

Based on the result shown in Figure 6, the loop height is determined to be H_para = 11 mm, corresponding to 0.09 wavelength at f_GLmax = 2.58 GHz. Figure 7 shows the frequency response of the gain. The bandwidth (BW) for a 3-dB gain drop criterion for G_L, denoted as G_L-BW, is 13.5% and the BW for a 3-dB gain drop criterion for G_R, denoted as G_R-BW, is 14.6%. For confirmation of the analysis/simulation results, measured/experimental results are also presented (see fabricated antenna in Fig. 5b).

Fig. 5 Metaloop with a parasitic natural conducting loop. (a) Perspective view. (b) Fabricated metaloop.

Fig. 6 GL at f = 2.58 GHz and GR at 3.51 GHz as a function of parasitic loop (paraloop) height Hpara, where wpara = 2 mm and Lpara = 116.7 mm (≈1λ0 at 2.58 GHz) are used.

Fig. 7 Frequency response of the gain with a parasitic loop (paraloop), where Hpara = 11 mm = 0.09 wavelength at fGLmax = 2.58 GHz.

5 Radiation pattern and VSWR

Figure 8a shows the analysis/simulation results of the radiation patterns at the maximum-gain low frequency f_GLmax = 2.58 GHz (radiation efficiency of η ≈ 69%) and high frequency f_GRmax = 3.51 GHz (η ≈ 60%), together with experimental results, where E_L and E_R denote the LHCP wave component and the RHCP wave component, respectively. For comparison, the analysis/simulation results in the absence of a parasitic loop are also shown in Figure 8b. It is clearly seen that the radiation pattern at low frequency f_GLmax is narrowed by virtue of the presence of the parasitic loop, while the radiation pattern at high frequency f_GRmax is less affected by the parasitic loop, as desired. Figure 9 shows the frequency response of the VSWR, which remains almost unchanged in the presence of the parasitic loop. Discrepancy between the analysis and experiment results is attributed to the fact that soldering the capacitive chips to the subwavelength segments is not uniform due to handwork.

Finally, the following comments are added. The metaloop antenna in this paper could be operated as a dual-band CP element with the same rotational sense (dual-band mono-CP radiation), although this is not our objective. Such dual-band mono-CP radiation is performed by introducing switching circuits to points F and T so that each point can be chosen to be either a feed point or a terminal point. (1) For dual-band mono-LHCP radiation, set points F and T to be the feed (fd) and terminal (trmnl) points, respectively, at low frequency f_low < f_T, which is expressed as . And at a high frequency f_high > f_T, change the role of points F and T by using switching circuits: . (2) For dual-band mono-RHCP radiation, set at f_low and at f_high by using switch-circuits.

Fig. 8 Normalized radiation patterns at fGLmax = 2.58 GHz and fGRmax = 3.51 GHz. (a) In the presence of a parasitic loop. (b) In the absence of a parasitic loop.

Fig. 9 Frequency response of the VSWRs in the presence and absence of a parasitic loop (paraloop). The 3-dB gain-drop bandwidths for GL and GR are denoted as GL-BW and GR-BW, respectively.

6 Conclusions

The dual-band counter CP wave radiated by a square metaloop antenna has a maximum gain of G_Lmax at frequency f_GLmax that is different from maximum gain G_Rmax at frequency f_GRmax, where G_Lmax < G_Rmax. To reduce the difference in these gains, a square parasitic natural conducting loop of one free-space wavelength at f_GLmax is placed at height H_para above the metaloop. It is found that there is an antenna height where the parasitic loop increases gain G_L, while not remarkably affecting gain G_R. Thus, the gain difference can be reduced, i.e., G_Lmax ≈ G_Rmax, with the VSWR remaining almost unchanged in the presence of the parasitic loop.

References

All Tables

All Figures

	Fig. 1 Natural loop antennas radiating a circularly polarized wave with a single feed point F. The loop has perturbation elements [2].
In the text

	Fig. 2 Metaloop antenna. (a) Perspective view. (b) Side view. (c) Arm unit cell.
In the text

	Fig. 3 Loop length L1 + L2 + L3 + L4 + L5 normalized to guided wavelength λg, as a function of frequency. β < 0 at f < fT, β = 0 at fT, and β > 0 at f > fT.
In the text

	Fig. 4 Frequency response of the gain.
In the text

	Fig. 5 Metaloop with a parasitic natural conducting loop. (a) Perspective view. (b) Fabricated metaloop.
In the text

	Fig. 6 GL at f = 2.58 GHz and GR at 3.51 GHz as a function of parasitic loop (paraloop) height Hpara, where wpara = 2 mm and Lpara = 116.7 mm (≈1λ0 at 2.58 GHz) are used.
In the text

	Fig. 7 Frequency response of the gain with a parasitic loop (paraloop), where Hpara = 11 mm = 0.09 wavelength at fGLmax = 2.58 GHz.
In the text

	Fig. 8 Normalized radiation patterns at fGLmax = 2.58 GHz and fGRmax = 3.51 GHz. (a) In the presence of a parasitic loop. (b) In the absence of a parasitic loop.
In the text

	Fig. 9 Frequency response of the VSWRs in the presence and absence of a parasitic loop (paraloop). The 3-dB gain-drop bandwidths for GL and GR are denoted as GL-BW and GR-BW, respectively.
In the text