Issue
EPJ Appl. Metamat.
Volume 6, 2019
Metamaterials Research and Development in Japan
Article Number 3
Number of page(s) 5
DOI https://doi.org/10.1051/epjam/2018010
Published online 18 January 2019

© H. Nakano et al., published by EDP Sciences, 2019

Licence Creative Commons
This is an Open Access article distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/4.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

1 Introduction

A loop antenna radiates a linearly polarized (LP) wave [1]. When the circumference of the loop is one wavelength, the maximum LP radiation appears in the broadside direction. This broadside LP radiation can be changed to circularly polarized (CP) radiation by adding perturbation elements [2] to the loop, as shown in Figure 1. The rotational sense of the CP radiation, either a right-handed (RH) sense or a left-handed (LH) sense, is uniquely determined by the location of the perturbation elements relative to feed point F: the loop in Figure 1 radiates an RHCP wave. In other words, the loop antenna radiates a CP wave with a single rotational sense.

The single rotational sense also holds true for a grid loop array in [3]. Although perturbation elements are not added to the loops, a traveling current is generated along the arrayed loops and radiates a CP wave. The rotational sense is determined by the winding sense of the loops.

It is often required that an antenna has dual-band counter CP radiation to keep sufficient separation of signals, i.e., the LHCP and RHCP radiation in two different frequency bands, in order to avoid interfering with each other. When a pair of dual-band counter CP antennas is adopted as a transmitting antenna and a receiving antenna, it is desirable for these antennas to have the same gain. If the LHCP and RHCP gains are different, the receiving antenna captures LHCP and RHCP waves with different receiving power levels. Therefore, a post-process circuit connected to the receiving antenna needs additional amplification circuits to enhance the weak power. This complicates the post-process circuit designs. To avoid such an issue, the same gain (balanced gain) is desired.

Recent study has shown that a metamaterial (MTM) loop antenna, referred to as a metaloop, is an antenna that meets the requirement of dual-band counter CP radiation [4], where the loop is realized using the concept of a composite RH and LH transmission line [57], referred to as a metaline.

The maximum gain for the metaloop within the low frequency band (LoFB) in [4] is smaller than that within the high frequency band (HiFB). This is attributed to the antenna size relative to the free-space wavelength (electrical antenna size); the free-space wavelength within the LoFB is larger than that within the HiFB, and hence the electrical antenna size within the LoFB is smaller than that within the HiFB.

Thus, a question arises as to how the difference in the gains can be reduced. This paper presents a technique for reducing the band gain difference for a dual-band counter CP metaloop antenna, where a parasitic loop (paraloop) is placed above the metaloop. Note that the metaloop antenna in this paper is analyzed using a full wave analysis tool, HFSS [8], and experimental work is performed in an anechoic chamber.

So far, other authors have designed MTM-loaded and MTM-inspired loop antennas. For example, the MTM-loaded loop in reference [9] and MTM-inspired loop [10] are designed as an LP antenna. An antenna using complementary capacitively loaded loop in reference [11] and loop antennas in [1220] are also designed as an LP antenna. The design requirements for these antennas differ from ours, i.e., dual-band counter CP radiation with balanced gain, while meeting two additional requirements: (1) broadside radiation and (2) simple feed system without balun circuits. To our best knowledge, there have not been such MTM-related loop antennas.

thumbnail Fig. 1

Natural loop antennas radiating a circularly polarized wave with a single feed point F. The loop has perturbation elements [2].

2 Metaloop structure

The metaloop antenna to be considered here is shown in Figure 2. The antenna arm is printed on a square dielectric substrate of side length Ssub, relative permittivity εr and thickness B. The substrate is backed by a square ground plane (GP) of side length SGP (=Ssub). The antenna arm, symmetric with respect to the xz plane, is composed of five straight metalines, whose lengths are L1 = L5 and L2 = L3 = L4. Each metaline is made of numerous conducting subwavelength segments of width w and length p0, where the gap between neighboring segments is denoted as Δg. Repeating arm sections, each having length 2(Δg + p0) ≡ p, are designated as the arm unit cells. The central segment of the cell is short-circuited to the GP through a chip inductor, LY. Neighboring segments are connected through a chip capacitor, 2CZ. Point F is the feed point and point T is the terminal point connected to the GP through a resistive load, RB, to suppress reflected currents from point T to point F.

Note that LY and 2CZ are determined as follows. First, using HFSS [8], we obtain the frequency response of the scattering parameters (S-parameters) for the unit cell including LY and 2CZ. Second, based on the obtained S-parameters, we draw the dispersion curve (phase constant β = ±2π/λg vs. frequency f [4], where λg is the guided wavelength). These two steps are repeated until the dispersion curve is balanced (smoothly connected at a preselected transition frequency fT), changing LY and 2CZ. The values for LY and 2CZ when the dispersion curve is balanced are what we need.

The parameters used for the following discussion are summarized in Table 1; these parameters create a preselected transition frequency of fT = 3 GHz, i.e., phase constant β is negative at frequencies below fT and positive at frequencies above fT. Figure 3 shows the loop peripheral length L1 + L2 + L3 + L4 + L5 normalized to the guided wave length, λg, as a function of frequency f. Note that the frequency-dependent guided wavelength λg used in Figure 3 is derived from the dispersion curve explained in the previous paragraph.

Frequencies fL and fU in Figure 3 are the lower- and upper-edge frequencies for a fast wave region. The radiation occurs between fL and fU. A traveling wave current flows along the loop from point F to point T, with guided wave length λg, which varies with frequency f. The propagation phase constant is β = −2π/λg < 0 at f < fT, β = 0 at f = fT, and β = 2π/λg > 0 at f > fT. In order to obtain a broadside beam at two different frequencies where the loop length is 1λg, the in-phase condition (β = 0) is not used.

thumbnail Fig. 2

Metaloop antenna. (a) Perspective view. (b) Side view. (c) Arm unit cell.

Table 1

Parameters.

thumbnail Fig. 3

Loop length L1 + L2 + L3 + L4 + L5 normalized to guided wavelength λg, as a function of frequency. β < 0 at f < fT, β = 0 at fT, and β > 0 at f > fT.

3 Frequency response of the gain

By resistive load RB, a traveling wave current flows along the loop from point F to point T, with frequency-dependent guided wavelength λg. In other words, the metaline acts as a leaky wave antenna. Generally, the radiation efficiency of the leaky wave antenna is not high due to absorption of the power input to the antenna by RB. If RB is removed, the input impedance (and hence Voltage Standing Wave Ratio (VSWR)) becomes unstable and CP radiation is not obtained by a reflected current from point T.

The current at frequency f < fT has a progressive phase distribution from point F to point T due to a negative phase constant (β = −2π/λg < 0). Hence, the current behaves as if it travels from point T to point F (clockwise). This results in LHCP radiation in the broadside direction at a frequency where the loop length is one guided wave length (1λg). Conversely, the current at f > fT has a regressive phase distribution from point F to point T due to a positive phase constant (β = 2π/λg > 0). Hence, the current flows from point F to point T (counter clockwise), resulting in RHCP radiation in the broadside direction at a frequency satisfying a 1λg-loop length. Thus, dual-band counter CP radiation in the broadside direction is obtained.

From Figure 3, it is expected that CP broadside radiation will be obtained around frequencies 2.60 GHz ≡ fN and 3.50 GHz ≡ fH, because the loop length normalized to guided wavelength λg is one at fN and fH: (L1 + L2 +…+L5)/λg = 1. For the remainder of the paper, fN and fH are referred to as the Nion frequency and Hion frequency, respectively. Figure 4 shows the gain as a function of frequency, where GL denotes the gain for an LHCP wave, called the LHCP gain, and GR denotes the gain for an RHCP wave, called the RHCP gain. It is found that GL is dominant at frequencies below fT = 3 GHz, because the current flows with a negative phase constant (β < 0). Conversely, GR is dominant at frequencies above transition frequency fT due to β > 0. Maximum gain GL appears at a frequency near Nion frequency fN and maximum gain GR appears at a frequency near Hion frequency fH. These frequencies are denoted as fGLmax and fGRmax, respectively. The difference in the maximum gains, GLmax at 2.58 GHz ≡ fGLmax (close to fN) and GRmax at 3.51 GHz ≡ fGRmax (close to fH), is approximately 5.5 dB.

thumbnail Fig. 4

Frequency response of the gain.

4 Reduction in the gain difference

In this section, the difference in maximum gains GLmax and GRmax is reduced as much as possible. For this, the antenna system shown in Figure 5 is considered, where a parasitic natural conducting strip loop of width wpara and peripheral length Lpara is located at height Hpara above the metaloop. The parasitic loop is designed such that only the smaller gain, GLmax at fGLmax, is increased, without reducing the larger gain, GRmax at fGRmax.

The CP radiation from the metaloop excites the parasitic loop and generates a rotating/traveling current along the loop.

The parasitic loop acts as a director for the driven meta loop, like Yagi–Uda antenna. If the radiation from the para sitic loop at fGLmax is constructively superimposed onto the radiation from the metaloop in the broadside direction, gain GLmax increases. For this to occur, parasitic loop length Lpara is chosen to be one free-space wavelength (1λ0) at fGLmax: Lpara = 116.7 mm. The remaining task is to optimize loop height Hpara.

Figure 6 shows GL at fGLmax = 2.58 GHz and GR at fGRmax = 3.51 GHz in the broadside direction (z-direction) as a function of parasitic loop height Hpara. It is found that there is an optimum antenna height for GLmaxGRmax.

Based on the result shown in Figure 6, the loop height is determined to be Hpara = 11 mm, corresponding to 0.09 wavelength at fGLmax = 2.58 GHz. Figure 7 shows the frequency response of the gain. The bandwidth (BW) for a 3-dB gain drop criterion for GL, denoted as GL-BW, is 13.5% and the BW for a 3-dB gain drop criterion for GR, denoted as GR-BW, is 14.6%. For confirmation of the analysis/simulation results, measured/experimental results are also presented (see fabricated antenna in Fig. 5b).

thumbnail Fig. 5

Metaloop with a parasitic natural conducting loop. (a) Perspective view. (b) Fabricated metaloop.

thumbnail Fig. 6

GL at f = 2.58 GHz and GR at 3.51 GHz as a function of parasitic loop (paraloop) height Hpara, where wpara = 2 mm and Lpara = 116.7 mm (≈1λ0 at 2.58 GHz) are used.

thumbnail Fig. 7

Frequency response of the gain with a parasitic loop (paraloop), where Hpara = 11 mm = 0.09 wavelength at fGLmax = 2.58 GHz.

5 Radiation pattern and VSWR

Figure 8a shows the analysis/simulation results of the radiation patterns at the maximum-gain low frequency fGLmax = 2.58 GHz (radiation efficiency of η ≈ 69%) and high frequency fGRmax = 3.51 GHz (η ≈ 60%), together with experimental results, where EL and ER denote the LHCP wave component and the RHCP wave component, respectively. For comparison, the analysis/simulation results in the absence of a parasitic loop are also shown in Figure 8b. It is clearly seen that the radiation pattern at low frequency fGLmax is narrowed by virtue of the presence of the parasitic loop, while the radiation pattern at high frequency fGRmax is less affected by the parasitic loop, as desired. Figure 9 shows the frequency response of the VSWR, which remains almost unchanged in the presence of the parasitic loop. Discrepancy between the analysis and experiment results is attributed to the fact that soldering the capacitive chips to the subwavelength segments is not uniform due to handwork.

Finally, the following comments are added. The metaloop antenna in this paper could be operated as a dual-band CP element with the same rotational sense (dual-band mono-CP radiation), although this is not our objective. Such dual-band mono-CP radiation is performed by introducing switching circuits to points F and T so that each point can be chosen to be either a feed point or a terminal point. (1) For dual-band mono-LHCP radiation, set points F and T to be the feed (fd) and terminal (trmnl) points, respectively, at low frequency flow < fT, which is expressed as . And at a high frequency fhigh > fT, change the role of points F and T by using switching circuits: . (2) For dual-band mono-RHCP radiation, set at flow and at fhigh by using switch-circuits.

thumbnail Fig. 8

Normalized radiation patterns at fGLmax = 2.58 GHz and fGRmax = 3.51 GHz. (a) In the presence of a parasitic loop. (b) In the absence of a parasitic loop.

thumbnail Fig. 9

Frequency response of the VSWRs in the presence and absence of a parasitic loop (paraloop). The 3-dB gain-drop bandwidths for GL and GR are denoted as GL-BW and GR-BW, respectively.

6 Conclusions

The dual-band counter CP wave radiated by a square metaloop antenna has a maximum gain of GLmax at frequency fGLmax that is different from maximum gain GRmax at frequency fGRmax, where GLmax < GRmax. To reduce the difference in these gains, a square parasitic natural conducting loop of one free-space wavelength at fGLmax is placed at height Hpara above the metaloop. It is found that there is an antenna height where the parasitic loop increases gain GL, while not remarkably affecting gain GR. Thus, the gain difference can be reduced, i.e., GLmaxGRmax, with the VSWR remaining almost unchanged in the presence of the parasitic loop.

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Cite this article as: Hisamatsu Nakano, Ittoku Yoshino, Tomoki Abe, Junji Yamauchi, Balanced gain for a square metaloop antenna, EPJ Appl. Metamat. 6, 3 (2019)

All Tables

Table 1

Parameters.

All Figures

thumbnail Fig. 1

Natural loop antennas radiating a circularly polarized wave with a single feed point F. The loop has perturbation elements [2].

In the text
thumbnail Fig. 2

Metaloop antenna. (a) Perspective view. (b) Side view. (c) Arm unit cell.

In the text
thumbnail Fig. 3

Loop length L1 + L2 + L3 + L4 + L5 normalized to guided wavelength λg, as a function of frequency. β < 0 at f < fT, β = 0 at fT, and β > 0 at f > fT.

In the text
thumbnail Fig. 4

Frequency response of the gain.

In the text
thumbnail Fig. 5

Metaloop with a parasitic natural conducting loop. (a) Perspective view. (b) Fabricated metaloop.

In the text
thumbnail Fig. 6

GL at f = 2.58 GHz and GR at 3.51 GHz as a function of parasitic loop (paraloop) height Hpara, where wpara = 2 mm and Lpara = 116.7 mm (≈1λ0 at 2.58 GHz) are used.

In the text
thumbnail Fig. 7

Frequency response of the gain with a parasitic loop (paraloop), where Hpara = 11 mm = 0.09 wavelength at fGLmax = 2.58 GHz.

In the text
thumbnail Fig. 8

Normalized radiation patterns at fGLmax = 2.58 GHz and fGRmax = 3.51 GHz. (a) In the presence of a parasitic loop. (b) In the absence of a parasitic loop.

In the text
thumbnail Fig. 9

Frequency response of the VSWRs in the presence and absence of a parasitic loop (paraloop). The 3-dB gain-drop bandwidths for GL and GR are denoted as GL-BW and GR-BW, respectively.

In the text

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